Electronic switch, and corresponding device and method

ABSTRACT

A high-voltage electronic switch includes first and second transistors defining a current flow path between an input and output of the switch. The transistors have a common point of the current flow path and a common control terminal. A control circuit includes a voltage line receiving a limit operating voltage and first and second branches coupled between the voltage line and the common point and common control terminal, respectively. Further transistors are activated, upon turning-off of the first and second transistors, for coupling the branches to the voltage line. The branches include a parallel connected resistor, diode, and string of diodes with opposite polarities. The diode of the first branch plus string of diodes of the second branch and diode of the second branch plus string of diodes of the first branch provide coupling paths between the voltage line and, respectively, the common point and common control terminal.

BACKGROUND Technical Field

The present disclosure relates to electronic switches. One or more embodiments regard high-voltage electronic switches that can be used, for example, in ultrasound-imaging applications.

Description of the Related Art

High-voltage switches are components that enable setting-up of a connection or isolation between different electrical paths within systems which use signals that may, for instance, be higher than 10 V and enable passage of currents that may, for example, be higher than 1 A.

The fields of application for these components are multiple.

For instance, the capacity to withstand high working voltages, at the same time maintaining the same electrical characteristics of a classic low-voltage switch, renders electronic switches components of election for systems, such as diagnostic imaging systems both in the medical field (e.g., ultrasonography machines), and in the industrial field (e.g., in the vehicle sector or for seeking micro-cracks within solid bodies). These components may, on the other hand, also be used in a wide range of consumer applications, such as to require management of power signals.

As an alternative to electronic switches, it is possible to use mechanical switches. These, however, present a rather high encumbrance (and hence a large occupation of space) and a lower reliability as compared to switches, in particular high-voltage switches, obtained with solid-state components of semiconductor material, for example, silicon.

Manufacture of these devices is facilitated by technologies, such as BCD (Bipolar-CMOS-DMOS) technologies, which enable production of high-voltage (HV) components and at the same time enable management of low-voltage (LV) digital signals for control of the state of the switches.

In particular, in the case where input or output voltages that are negative with respect to ground, production of these devices is facilitated by the use of HV Sol (Silicon-on-Insulator) technologies in order to avoid the presence of potentially harmful parasitic components between the various switches.

BRIEF SUMMARY

One or more embodiments provide further improvements in the sector of electronic switches.

One or more embodiments regard a corresponding device (e.g., a switching device of the same type as those recalled in the introductory part of the present description) and a corresponding method.

The claims form an integral part of the technical teaching provided herein in relation to the embodiments.

One or more embodiments enable optimization of the step of turning-off of the switch by exploiting the positive and negative transitions of the input and output signals.

One or more embodiments enable increase of isolation as the slew rate of the signals applied increases, thus increasing isolation of the switch in conditions of maximum criticality.

The structure that can be obtained according to one or more embodiments does not have nodes sensitive to parasitic capacitances, there being, instead, the possibility of exploiting these factors in order to optimize operation of the switch, rendering the step of design and layout of the device simpler and without any risk of failure, associated, for example, to the off-state-driving block.

In one or more embodiments, driving of the control terminal (e.g., the gate) of the high-voltage devices does not require complex structures, it being possible for the control signal to can be obtained through current mirrors, without generating negative (and positive) reference voltages.

In one or more embodiments, the power consumed in the off state of the switch is practically zero.

One or more embodiments enable operation of each component within the voltage class for which it has been designed, eliminating the conditions of breakdown or deterioration over time.

One or more embodiments enable turning-off to be rendered effective for any value of the input voltage V_(IN) comprised between supply values VPP and VNN.

In one or more embodiments, the turning-off condition is independent of the HV supply voltages used for supplying the driving structures of the switch (e.g., V_(GS)).

One or more embodiments enable creation of a structure that is not liable to create harmonic distortion on the signals when the switch is on.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

One or more embodiments will now be described, purely by way of non-limiting example, with reference to the annexed drawings, wherein:

FIG. 1 is a circuit diagram exemplifying one or more embodiments; and

FIG. 2 exemplifies possible timing charts of signals in some embodiments.

DETAILED DESCRIPTION

In the ensuing description various specific details are illustrated and aimed at providing an in-depth understanding of examples of one or more embodiments. The embodiments may be obtained without one or more of the specific details, or with other methods, components, materials, etc. In other cases, known structures, materials, or operations are not illustrated or described in detail so that various aspects of the embodiments will not be obscured. Reference to “an embodiment” or “one embodiment” in the framework of the present description is intended to indicate that a particular configuration, structure, or characteristic described in relation to the embodiment is comprised in at least one embodiment. Hence, phrases such as “in an embodiment” or “in one embodiment” that may be present in various points of this description do not necessarily refer to one and the same embodiment.

The different topologies used for production of electronic switches, for example, of the high-voltage (HV) type, may depend upon the electrical characteristics that it is intended to obtain and upon the sphere of application for which they are designed.

As exemplified in FIG. 1, a structure that can be used to produce a switch 10 (e.g., an HV switch) may envisage use of two transistors HVMN1 and HVMN2 (e.g., N-channel power DMOS transistors).

By connecting together the current-source terminals S of the transistors HVMN1 and HVMN2 and doing the same also for the (gate) control terminals G, it is possible to obtain two common terminals GATE 12 and SOURCE 14 for driving simultaneously both of the devices HVMN1 and HVMN2, which are arranged in cascaded fashion (i.e., with the respective current paths, for example, drain-to-source and source-to-drain paths, in series with one another) between two terminals, an input terminal IN and an output terminal OUT, of the switch 10.

In this way, the current flow path between the terminals IN and OUT may be conductive (on) or non-conductive (off) according to whether the two transistors HVMN1 and HVMN2 are conductive or non-conductive.

The control terminal 12 may be driven by a driver module (not visible in the figures), provided in the device in which the switch is mounted. The corresponding driving action may be performed according to known criteria, which renders superfluous a more detailed description herein.

The main advantage of these devices is represented by the high switching rate and by the good efficiency also at low power levels. The high breakdown voltage and the low resistance in triode conditions R_(ON) renders the component ideal for providing integrated HV switches, especially within low-power-consumption systems.

Proper operation of the switch is facilitated by the capacity of the transistors (e.g., HVMN1 and HVMN2) to withstand high voltage levels. For example, in the case where the switch 10 is off, the value of voltage applied at the input may differ from the output by the maximum peak-to-peak voltage (V_(IN) _(PK-PK) ) of the input signal.

Any possible breakdown of the device may be prevented by getting the maximum voltage V_(IN) _(PK-PK) to coincide with or be lower than the voltage class of the HV MOSFETs used.

Considering the voltages involved and the interest in having a low resistance R_(ON) in the case where the switch is on, the MOS transistors used may be of large size so as to be able to withstand high signal currents, remaining in the triode zone. For example, N-channel HV MOSFETs may represent a good choice since they enable all the electrical characteristics just described to be achieved, occupying a smaller area on silicon as compared to P-channel HV MOSFETs.

A structure of the type described above may be driven by circuit blocks, which control the voltages at the common terminals GATE and SOURCE, enabling proper biasing of the state.

The on state of the switch can be established by bringing the gate voltage of the two HV MOSFETs to a value higher than the source voltage by at least a threshold voltage

V _(GS) _(ON) →V _(GS) >V _(TH)

where V_(GS)=V_(G)−V_(S), and V_(TH) is the threshold voltage.

Since the source forms part of the path between the input terminal IN and the output terminal OUT of the switch, this node may follow the input signal HV and be able to follow all the entire input dynamics. Likewise, also the gate node may be able to fluctuate freely so as to be able to maintain the correct on-state gate-to-source voltage V_(GS) _(ON) . This effect can be obtained by charging the parasitic capacitance C_(GS) (represented dashed) between the two terminals GATE and SOURCE. This capacitance is mainly made up of the parasitic capacitances of the MOSFETs C_(GS) _(HVMN1) and C_(GS) _(HVMN2) , which help to keep the state stored.

The off condition of the switch 10 may be obtained by driving the two nodes GATE and SOURCE in order to obtain

V _(GS) _(OFF) →V _(GS)≦0V

A good off-state-driving circuit enables maintenance of the off state in a stable and constant way even in the presence of input signals with high voltage and high slew rate SR. This condition may be important above all in the presence of high-voltage negative input signals.

To achieve the off condition of the switch, it is possible to resort to various circuit solutions.

A first solution, as described, for example, in U.S. 2012/0249261 A1, consists in pre-charging of the capacitance C_(GS) in an opposite way with respect to the on condition, i.e., setting the source at a voltage higher than the gate by a value that satisfies the relation V_(G)≦V_(S)≦V_(G)+|V_(GS) _(MAX) |, where |V_(GS) _(MAX) | represents the highest value of |V_(GS)| (in modulus) allowed by the technology without breakdown or deterioration of the component. During the switchings of V_(IN), both of the nodes remain disconnected from any reference in voltage and can hence float following upon sudden variations of the signals across the switch.

The main disadvantages of this solution are due to the negative impact that the parasitic capacitances in the gate and source nodes have in keeping stable the off state of the switch, to the worsening of the harmonic distortion due to the modulation of the gate-to-source voltage V_(GS), and to the critical working conditions of some high-voltage components, which may undergo very repetitive electronic stresses that may even cause breakdown of the component.

Also to be taken into account is the need to know the biasing beforehand so as to be able to pre-charge the voltage V_(GS) correctly. This condition is not always satisfied in applications in general.

Typically, the values allowed for the signal at input to the switch cannot exceed the HV voltages used within the driver circuits.

A second solution, which is described, for example, in S-J Jung et al.: “Area-efficient high-voltage switch using floating control circuit for 3D ultrasound imaging systems”—ELECTRONICS LETTERS, Dec. 4, 2014, Vol. 50, No. 25, pp-1900-1902 consists in forcing the gate and source nodes to one or more reference voltages (typically the lowest of the device). In the case where both of the nodes are biased at the most negative voltage of the circuit (VNN), turning-off of the switch may be obtained with

V _(G) =V _(S) =VNN→V _(GS)=0V

The voltage VNN reverse biases the body-drain diodes of the transistors HVMN1 and HVMN2, ensuring constant and independent isolation from the voltage present at input to the circuit.

The main disadvantage of this solution is represented by the considerable sensitivity in maintaining the off state following upon fast edges of the input signal.

A possible variant of this solution consists in biasing the source node at VNN and the gate at a voltage higher than V_(S)=VNN

VNN<V _(G) ≦VNN+|V _(GS) _(MAX) |

thus obtaining an off state that is more robust in regard to couplings with the input signal thanks to a negative gate-to-source voltage V_(GS)

0<V _(GS) ≦−|V _(GS) _(MAX) |

This technique in any case does not altogether solve coupling and requires protection circuits in order not to stress the components used for generation of the references.

These two solutions may prove more robust to the parasitic capacitances as compared to the solution described previously.

Finally, for some topologies, stable maintenance of the state of the switch may imply a constant consumption from the HV supplies, which increases the power dissipated by the circuit as a whole.

One or more embodiments enable solution of the problems outlined above, for example in circuits with negative input voltages.

One or more embodiments may envisage recourse to an off-state-driving block according to the solution exemplified in FIG. 1.

In one or more embodiments, the control circuit in question may use two further transistors (e.g., two high-voltage MOSFETs HVMN3 and HVMN4), driven by a common signal RST applied to the (gate) control terminals of the transistors HVMN3 and HVMN4, so as to be able to connect two nodes A and B (e.g., on the drains of HVMN3 and HVMN4) to a voltage line 16 that is to be coupled to a source (comprised in the device in which the switch is mounted and not visible in the figures), which supplies a limit operating voltage. This may, for example, be a negative voltage, which may correspond to the voltage VNN, i.e., to the lowest voltage of the circuit, already referred to previously.

In one or more embodiments, the signal RST can be generated via a driver module P, which is in itself known, this rendering superfluous a more detailed description herein. The module P may be able to activate the further transistors HVMN3, HVMN4 upon turning-off of the “main” transistors HVMN1 and HVMN2 in order to couple the respective branch 1 a, 1 b of the control circuit to the voltage line 16.

In one or more embodiments, the transistor HVMN3 can be coupled to the terminal or point SOURCE (designated by 14) via a first circuit branch 1 a, which may comprise a connection in parallel of:

-   -   a resistance R_(S);     -   a diode (e.g., a low-voltage diode) D_(S0) connected to HVMN3         (node A) via the anode, whereas the cathode is connected to the         node S, for example, via the source; and     -   a string of N diodes (N>1) connected together in series and with         their cathodes set facing the node A.

In one or more embodiments, the transistor HVMN4 may be coupled to the branch GATE (designated by 12) via a second circuit branch 1 b, which may comprise a connection in parallel of:

-   -   a resistance R_(G);     -   a diode D_(G0) with its cathode connected to the node B; and     -   a string of M diodes connected together in series and with their         cathodes se facing the control (gate) terminals of the two         “main” transistors HVMN1 and HVMN2 that form the switch 10         according to the criteria exemplified previously.

In one or more embodiments, the circuit exemplified in FIG. 1 may operate according to the modalities described hereinafter.

During the on state (switch conductive), the signal RST (which may be a logic signal, e.g., high/low) referenced around the value VNN goes to a low level (e.g., VNN), keeping the transistors HVMN3 and HVMN4 off.

In this way, the nodes A and B can follow the entire dynamics of the input signal applied, with a voltage class of the HV MOSFETs HVMN1 and HVMN2 that is equal to or higher than (VPP+|VNN|).

During the off state, the signal RST may be at a high level so as to turn on the transistors HVMN3 and HVMN4, in order to bring the nodes A and B to the voltage VNN, in d.c. conditions, with the two resistances R_(S) and R_(G) that are such as to achieve

V _(G) =V _(S) =VNN→V _(GS)=0V

One or more embodiments may intrinsically facilitate regeneration of the off state, with an efficiency proportional to the slew rate of the input signal.

The chronograms of FIG. 2 exemplify conditions of operation where there is applied on the input IN a square wave of amplitude comprised between VNN and VPP, where VPP is the highest HV voltage that it is assumed that the switch 10 is able to withstand. It may, for example, be the HV supply of the on-state-driving block.

Following upon a transition of the input signal, the drain-to-source (C_(IN-SOURCE)) and drain-to-gate (C_(IN-GATE)) parasitic capacitances of the transistor HVMN1 (see FIG. 1) operate in such a way as to cause the nodes GATE 12 and SOURCE 14 to follow in a.c. conditions the input node IN (but for a capacitive partition). In itself, such a variation of voltage on the two control nodes is not easy to handle, in so far as it may depend both upon the parasitic structures of the components used and upon the ones introduced in the stage of layout of the device. This effect may produce a variation of the voltage V_(GS), with the possibility of obtaining an undesired change of state of the switch 10 following upon a transition of the input signal. In general, a reduction of the input voltage corresponds to a drop in the voltages V_(GATE) and V_(SOURCE) and vice versa.

In one or more embodiments, this problem can be tackled with the circuits 1 a and 1 b.

During the positive transition of the input signal at time t₀ (see FIG. 2) both the voltage on the gates of the two transistors HVMN1, HVMN2 and the voltage on their sources undergo an increase in their nominal value. This variation forward biases (direct polarization) the diode D_(G0) and the string of diodes D_(S1-SN), clamping the variation of voltage of the two nodes to the values

V_(GATE↑) = VNN + V_(DS_(HVMN 34)) + V_(D_(G0)) $V_{{SOURCE}\; \uparrow} = {{VNN} + V_{{DS}_{{HVMN}\; 34}} + {\sum\limits_{x = 1}^{N}\; V_{D_{Sx}}}}$ with  V_(DS_(HVMN 3)) = V_(DS_(HVMN 4)) = V_(DS_(HVMN 34)).

Considering now the instant at which the input signal produces a negative transition (instant t₁ in FIG. 2), the nodes GATE 12 and SOURCE 14 decrease until they cause forward biasing of the body-to-drain diodes of the transistors HVMN3 and HVMN4, the diode D_(S0), and the string of diodes D_(G1-GN), with the two voltages hence clamped at the values

V_(SOURCE ↓) = VNN − V_(D_(HVMN 34)) − V_(D_(S 0)) $V_{{GATE} \downarrow} = {{VNN} - V_{D_{{HVMN}\; 34}} + {\sum\limits_{x = 1}^{N}\; V_{D_{Gx}}}}$

In both cases, the above effect intrinsically implies V_(G)<V_(S). Moreover, by confining all the nodes of the structure to a minimum value of

${VNN} - V_{D_{{HVMN}\; 34}} + {\sum\limits_{x = 1}^{N}\; V_{D_{G\; x}}}$

it is possible to obtain an adequate safety for the components used.

Thanks to the limitations in voltage that may be introduced in one or more embodiments, the control voltage of the switch not only does not change the state of the switch, but further reduces the voltage V_(GS) to values lower than 0 V, facilitating making turning-off more robust in respect of the d.c. turning-off value.

It will be appreciated that the same principle applies also considering the effect of an external HV signal applied to the output node OUT of the switch 10.

One or more embodiments may draw benefit from the fact that the “parasitic” capacitances produced by the transistors HVMN3 and HVMN4 on the nodes A and B may be rendered substantially identical. This result may be achieved by providing the two components with substantially the same area

W_(HVSW1)=W_(HVSW2)

L_(HVSW1)=L_(HVSW2)

In this way, it is possible to make it easier to obtain the same current capacity and a consequent symmetry of the effects both on the node GATE 12 and on the node SOURCE 14.

Moreover, in one or more embodiments, the parasitic capacitances C_(PAR) _(_) _(A), C_(PAR) _(_) _(B) and the capacitances C_(PAR) _(_) _(A) _(_) _(GD) and C_(PAR) _(_) _(B) _(_) _(GD) (see FIG. 1), amplified by the Miller effect on the nodes A and B, make it possible not only to get the circuits 1 a and 1 b to intervene more rapidly, but also to protect the low-voltage control signal RST by undesired voltage peaks.

One or more embodiments may involve a sizing of the two transistors HVMN1, HVMN2 founded, for example, on a synthesis between area occupation, maximum turning-off time, and robustness to transitions of the high-voltage signals across the switch.

The choice of the values that can be attributed to the resistances R_(S) and R_(G) can affect the performance of the turning-off driver in two different temporal phases.

For instance, a first phase may be at the transition of state from on to off. In fact, when the MOSFETs HVMN3 and HVMN4 have just turned on, they start to discharge the parasitic capacitances C_(PAR) _(_) _(A) and C_(PAR) _(_) _(B), and the current starts to flow in the two resistances, the voltage across the structures 1 a and 1 b being initially zero. Since the current of the resistances is inversely proportional to the value of R_(S) and R_(G), the voltage at the control nodes decreases in a time depending upon the two passive components. For instance, in the condition R_(S)=R_(G), the charge associated to the nodes SOURCE 14 and GATE 12 can decrease in the same way, initially keeping the value of V_(GS) constant in conditions of switch in the on state. By choosing instead R_(S)>R_(G), it is possible to cause the gate voltages of the transistors HVMN1 and HVMN2 to decrease more rapidly than the source voltages, accelerating turning-off of the switch. Once the voltage at the node A drops with respect to that on the node SOURCE by

$\sum\limits_{x = 1}^{N}\; V_{D_{Sx}}$

and the voltage at the node B drops with respect to that on the node GATE by V_(D) _(GD) , the discharge current starts to pass through the diodes, reducing V_(S) and V_(G) with a current equal to the maximum current capacity of the turning-off HV MOSFETs.

A second phase may, instead, arise in conditions of the switch 10 in the off state (V_(G)=V_(S)=VNN) and in concomitance with a variation of the input signal. Once the diodes of the circuits 1 a and 1 b limit the variation of the nodes SOURCE 14 and GATE 12 to the values described previously, the resistances R_(S) and R_(G) start to bring the control voltages of the switch back to the initial values. Consequently, the choice of the values to be attributed to the passive components may make it possible to be set up, together with the parasitic capacitances C_(GATE) and C_(SOURCE), the discharge time of the two nodes. The choice of a not excessively low value of these capacitances enables maintenance of a negative control voltage for the longest time possible. For instance, in one or more embodiments, it is possible to choose these values so as to privilege maintenance of the voltage V_(GS) _(OFF) during the most critical transition of the input signal. Assuming, as in FIG. 2, that the critical transition is in concomitance with the positive edge of the input signal, it is possible to choose R_(S)>R_(G) so as to maintain the control voltage negative for a longer time given that discharge of the voltage GATE is faster than that of the voltage SOURCE.

It will be appreciated that one or more embodiments may envisage (purely by way of illustrative and non-limiting example):

-   -   the possible choice of polarities of components and/or of         signals in time (e.g., positive/negative) complementary to those         presented herein by way of example, with consequent possible         adoption both of polarities of connection (e.g., location of         anode and cathode) and of polarities of components (e.g., N type         or P type) complementary, accordingly, to the ones exemplified         herein; and/or     -   the possible replacement of transistors exemplified herein as         field-effect transistors (e.g., MOSFETs) with bipolar         transistors or other similar components, where:     -   the current path between the current-source terminal and the         current-drain terminal, represented in a FET by the         source-to-drain path, is represented by emitter-collector path,         and/or     -   the control terminal, represented in a FET by the gate, is         represented by the base.

One or more embodiments may consequently provide an electronic switch that can be operated with a limit operating voltage (e.g., a negative voltage, VNN), the switch comprising:

-   -   a first transistor and a second transistor (e.g., HVMN1 and         HVMN2) arranged in cascaded fashion so as to provide a current         flow path between an input terminal and an output terminal of         the switch, the first and second transistors having between them         a common point (e.g., S, 14) of the current flow path and a         common control terminal (e.g., G, 12) for rendering the current         flow path conductive and non-conductive by turning on and         turning off the first and second transistors;     -   a control circuit, comprising a voltage line (e.g., 16), which         can be coupled to said limit operating voltage (e.g., VNN), and         comprising a first branch (e.g., 1 a) between said voltage line         and said common point and a second branch (e.g., 1 b) between         said voltage line and said common control terminal, wherein each         said first and second branch:     -   a) is coupled to a further transistor (e.g., HVMN3, HVMN4) that         can be activated, upon turning-off of the first and second         transistors, for coupling the respective branch of the control         circuit to said voltage line; and     -   b) comprises parallel connection of a resistor (e.g., R_(S),         R_(G)), a diode (e.g., D_(S0); D_(G0)), and a string of diodes         (e.g., D_(S1-SN); D_(G1-GN)), with said diode and said string of         diodes that have opposite polarities (i.e., position of their         anode with respect to their cathode), so that the diode in the         first branch plus the string of diodes in the second branch and         the diode in the second branch plus the string of diodes in the         first branch provide forward-biasing (direct polarization)         coupling paths (with the anode at a higher voltage than the         cathode) between said voltage line and, respectively, said         common point and said common control terminal.

In one or more embodiments, the resistors may have one and the same value for the first branch and for the second branch.

In one or more embodiments, the resistor in the first branch may have a higher value than the resistor in the second branch.

In one or more embodiments, the further transistors may substantially have the same area for the first branch and for the second branch.

In one or more embodiments, the first transistor and the second transistor have current-source terminals (e.g., source S) coupled together at the common point.

In one or more embodiments:

-   -   the diode (D_(S0)) in the first branch (1 a) and the diode         (D_(G0)) in the second branch (1 b) can have their cathodes set         facing, respectively, the common point and the voltage line; and     -   the diodes in the string of diodes in the first branch and the         diodes in the string of diodes in the second branch can have         their cathodes set facing, respectively, the voltage line and         the common point.

In one or more embodiments, the transistors may be field-effect transistors, optionally N-channel MOSFETs.

In one or more embodiments, a device comprising an electronic switch of the type exemplified herein that is able to provide a current flow path in the device may comprise:

-   -   a source of the limit operating voltage (e.g., VNN), which is         preferably negative, coupled to the voltage line; and     -   a driver device (e.g., the module P) for activating the further         transistors in the first and second branches upon turning-off of         the first and second transistors in order to couple the         respective branch (1 a, 1 b) of the control circuit to the         voltage line (16).

In one or more embodiments, a method of operation of an electronic switch of the type exemplified herein may envisage activation of the further transistors coupled to the first and second branches by coupling the respective branch of the control circuit to the voltage line upon turning-off of the first and second transistors.

Without prejudice to the underlying principle, the details of construction and the embodiments may vary, even significantly, with respect to what has been illustrated herein purely by way of non-limiting example, without thereby departing from the extent of protection of the present invention, which is defined by the annexed claims.

The various embodiments described above can be combined to provide further embodiments. These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure. 

1. An electronic switch, comprising: a first transistor and a second transistor, each transistor having a control node and a signal node, the signal nodes of the first and second transistors coupled in series between an input node and an output node and having a common signal node defined at an interconnection of the signal nodes of the first and second transistor, and the control nodes of the first and second transistors coupled to a common control node configured to receive a control signal to turn the first and second transistors on and off; and an off-state control circuit configured to provide a respective electrical path between a reference supply voltage node configured to receive a limit operating voltage and each of the common control node and the common signal node, the respective electrical paths configured, responsive to the control signal turning off the first and second transistors, to limit a voltage on the common control node relative to a voltage on the common signal node to keep the first and second transistors turned off in the presence of a capacitive coupling of a signal on the input node to the common control and common signal nodes.
 2. The electronic switch of claim 1, wherein the respective electrical paths are further configured to limit the voltage on the common control node relative to the voltage on the common signal node so the first and second transistors do not turn on in response to capacitive coupling of a signal on the output node to the common control and common signal nodes.
 3. The electronic switch of claim 2, wherein the electrical path between the common signal node and the reference supply voltage node comprises a first branch circuit coupled between the common signal node and a first branch node configured to be coupled to the reference supply voltage node, the first branch circuit including a resistive element coupled between the common signal node and the first branch node, a first diode element coupled between the common signal node and the first branch node to provide a forward-biased coupling path between the first branch node and the common node, and a first plurality of series-connected diode elements coupled between the common signal node and the first branch node to provide a forward-biased coupling path between the common node and the first branch node.
 4. The electronic switch of claim 3, wherein the first branch circuit comprises a first further transistor configured to couple the first branch node to the reference supply voltage node in response to the control signal having a value to turn off the first and second transistors.
 5. The electronic switch of claim 4, wherein the electrical path between the common control node and the reference supply voltage node comprises a second branch circuit coupled between the common control node and a second branch node configured to be coupled to the reference supply voltage node, the second branch circuit including a resistive element coupled between the common control node and the second branch node, a second diode element coupled between the common control node and the second branch node to provide a forward-biased coupling path between the common control node and the second branch node, and a second plurality of series-connected diode elements coupled between the common control node and the second branch node to provide a forward-biased coupling path between the second branch node and the common control node.
 6. The electronic switch of claim 5, wherein the second branch circuit comprises a second further transistor configured to couple the second branch node to the reference supply voltage node in response to the control signal having a value to turn off the first and second transistors.
 7. The electronic switch of claim 6, wherein the resistor element in each of the first and second branch circuits has a same value.
 8. The electronic switch of claim 7, wherein the first and second further transistors are substantially the same size transistor.
 9. The electronic switch of claim 1, wherein each of the first and second transistors comprises a field effect transistor.
 10. The electronic switch of claim 9, wherein each of the field effect transistors comprises an N-channel MOSFET.
 11. An electronic switch, comprising: a first transistor and a second transistor, each transistor having a control node and signal nodes, the signal nodes of the first and second transistors coupled in series between an input node and an output node and having a common node defined at an interconnection of the signal nodes of the first and second transistor, and the control nodes of the first and second transistors being coupled to a common control node; a control circuit including a reference supply voltage node configured to receive a limit operating voltage, the control circuit including: a first branch circuit coupled between the common node and a first branch node configured to be coupled to the reference supply voltage node, the first branch circuit including a resistive element coupled between the common node and the first branch node, a first diode element coupled between the common node and the first branch node to provide a forward-biased coupling path between the first branch node and the common node, and a first plurality of series-connected diode elements coupled between the common node and the first branch node to provide a forward-biased coupling path between the common node and the first branch node; and a second branch coupled between the common control node and a second branch node configured to be coupled to the reference supply voltage node, the second branch circuit including a resistive element coupled between the common control node and the second branch node, a second diode element coupled between the common control node and the second branch node to provide a forward-biased coupling path between the common control node and the second branch node, and a second plurality of series-connected diode elements coupled between the common control node and the second branch node to provide a forward-biased coupling path between the second branch node and the common control node.
 12. The electronic switch of claim 11 further comprising a third transistor and a fourth transistor configured to couple the first branch node and second branch node to the reference supply voltage node, respectively, in response to a reset signal.
 13. The electronic switch of claim 11 further comprising one of an ultrasound imaging system and an industrial imaging system for providing images showing micro-cracks in solid bodies.
 14. The electronic switch of claim 9, wherein each of the first and second transistors comprises either a field effect transistor or a bipolar transistor.
 15. The electronic switch of claim 1, wherein: the first diode element in the first branch circuit and the second diode element in the second branch circuit have cathodes coupled to the reference supply voltage node and the common control node, respectively; the first plurality of series-connected diode elements has a first diode having an anode coupled to the common signal node and a cathode of a last diode coupled to the first branch node; and the second plurality of series-connected diode elements has a first diode having an anode coupled to the second branch node and the cathode of a last diode coupled to the common control node.
 16. A method, comprising: turning off first and second switches coupled in series between an input node and an output node of the electronic switch, a common signal node defined at an interconnection of the first and second switches a common control node defined at an interconnection of control nodes of the first and second switches; limiting a value of a voltage on a common control node based upon the first and second switches being turned off; and limiting a value of a voltage on the common signal node based upon the first and second switches being turned off, the voltages on the common control and signal nodes being limited to values to so the first and second transistors remain turned off even when signals on the input node or output node are capacitively coupled to the common control and signal nodes.
 17. The method of claim 16, wherein limiting the value of the voltage on the common control node and the common signal node comprises coupling at least one rectification element between each of the common control node and the common signal node and a limit voltage node receiving a limit voltage.
 18. The method of claim 17, wherein coupling the at least one rectification element between each of the common control node and the common signal node and a limit voltage node receiving a limit voltage comprises coupling the at least one rectification element between each of the common control node and the common signal node and a limit voltage node receiving a limit voltage in response to turning off the first and second transistors.
 19. The method of claim 18, wherein turning off the first and second transistors comprises driving the common control node with a control signal having a value to turn off the first and second switches.
 20. The method of claim 17 further comprising supplying a negative reference voltage as the limit voltage on the limit voltage node. 